The output of a transmitter contains a desired transmit signal as well as Intermodulation Distortion (IMD) and thermal noise. The IMD is caused by non-linear behavior of components in the transmitter such as, for example, a power amplifier. Various linearization techniques (e.g., digital predistortion) are utilized to minimize IMD and, in some cases, thermal noise. However, even after linearization, there is still some thermal noise and residual IMD in the output of the transmitter. In the case of a frequency division duplex system, some of the thermal noise and the IMD exists in a paired receive band (i.e., in a receive band of a co-located receiver). Specifically, FIG. 1 illustrates a frequency domain representation of a concurrent dual-carrier transmit signal where some type of linearization has been applied to the transmit band. The paired uplink channel in the receive band can be seen to be occupied by residual IMD and thermal noise that is generated in the transmitter.
In order to address this issue, currently, a transmit band filter at the output of the transmitter is specified to exhibit an adequately deep stop-band in the paired receive band. This deep stop-band reduces the amount of thermal noise and residual IMD that leaks from the output of the transmitter into the co-located receiver. The deep stop-band requirements of the transmit band filter result in several issues. Specifically, the requirement for the transmit band filter to exhibit a deep stop-band increases the number of resonators needed for the transmit band filter and increases the time needed to tune the transmit band filter. Further, increasing the number of resonators increases a size of the transmit band filter, increases an insertion loss of the transmit band filter, and makes the transmit band filter more costly to manufacture. As such, it is desirable to relax the deep stop-band requirements of the transmit band filter.
One technique that has been used to relax the deep stop-band requirements of the transmit band filter is active cancellation of transmitter noise from the input of the co-located receiver. As used herein, “transmitter noise” includes both residual IMD after any linearization and thermal noise. More specifically, FIG. 2 illustrates a communication node 10 including a transmitter 12 and a receiver 14 in which a feedforward architecture is utilized to suppress or cancel transmitter noise in a receive band of the receiver 14. The receiver 14 is referred to herein as being co-located with the transmitter 12. As used herein, the term “co-located” is used to indicate that a transmitter and a receiver are both located at, or included as part of, a single communication node.
As illustrated, the transmitter 12 includes a Digital-to-Analog Converter (DAC) 16, an upconversion subsystem 18, and a power amplifier (PA) 20 connected as shown. The DAC 16 converts a digital baseband transmit signal (STX) into an analog baseband transmit signal (STX,AG). The upconversion subsystem 18 upconverts the analog baseband transmit signal (STX,AG) to a desired radio frequency to provide an upconverted transmit signal (STX,UP). The power amplifier 20 amplifies the upconverted transmit signal (STX,UP) to thereby provide an analog radio frequency transmit signal (STX,RF) at an output of the transmitter 12. The analog radio frequency transmit signal (STX,RF) is provided to an antenna 22 of the communication node 10 via a duplexer 24.
The receiver 14 includes a Low Noise Amplifier (LNA) 26, a downconversion subsystem 28, and an Analog-to-Digital Converter (ADC) 30 connected as shown. The LNA 26 amplifies an analog radio frequency receive signal (SRX,RF) received from the antenna 22 via the duplexer 24. A resulting amplified radio frequency receive signal (SRX,AMP) is downconverted to baseband via the downconversion subsystem 28 to thereby provide an analog baseband receive signal (SRX,AG). The analog baseband receive signal (SRX,AG) is digitized by the ADC 30 to provide a digital baseband receive signal (SRX) at an output of the receiver 14.
The analog radio frequency transmit signal (STX,RF) includes both a desired transmit signal in a transmit band of the transmitter 12 and transmitter noise. The transmitter noise includes thermal noise and IMD in a receive band of the receiver 14. The transmitter noise in the receive band of the receiver 14 leaks into the receiver 14 through the duplexer 24. In order to suppress or cancel the transmitter noise in the receive band of the receiver 14, the communication node 10 includes a feedforward transmit (TX) noise cancellation subsystem 32. As described in A. Roussel, C. W. Nicholls, and J. S. Wight, “Frequency agile bandstop filter (FABSF),” IEEE MTT-S International, pp. 1099-1102, June 2008 (hereinafter the “Roussel article”), the feedforward TX noise cancellation subsystem 32 includes a signal cancellation loop and an error cancellation loop. The signal cancellation loop is formed by couplers 34, 36, and 38, a complex gain element 40 (e.g., a Radio Frequency (RF) vector modulator), and a fixed delay line 42 connected as shown. The complex gain element 40 is tuned such that the signal cancellation loop cancels the desired signal from the analog radio frequency transmit signal (SRX,RF) at the coupler 38 to thereby provide a signal that is representative of the transmitter noise to the error cancellation loop.
The error cancellation loop is formed by a complex gain element 44 (e.g., an RF vector modulator), an error amplifier 46, a coupler 48, and a fixed delay line 50 connected as shown. In the error cancellation loop, the signal output by the signal cancellation loop is adjusted by the complex gain element 44 and then recombined with the analog radio frequency transmit signal (STX,RF) at the coupler 48. The complex gain element 44 is tuned to cancel the transmitter noise in the receive band of the receiver 14. The fixed delay lines 42 and 50 are utilized to minimize a group delay mismatch between the two paths (i.e., the feedforward path and the main path).
Simulation results show that the feedforward TX noise cancellation subsystem 32 described in the Roussel article could cancel the transmit noise in the receive band by around 30 decibels (dB), but only over a 5 Megahertz (MHz) bandwidth. As such, the feedforward TX noise cancellation subsystem 32 is not suitable for wideband or multiband applications such as, for example, Long Term Evolution (LTE) cellular communications networks. More specifically, the complex gain elements 40 and 44 use phase shifters or vector modulators. Phase shifters and vector modulators are limited in bandwidth and, as a result, limit the bandwidth of the feedforward TX noise cancellation subsystem 32. In addition to being limited in bandwidth, the feedforward TX noise cancellation subsystem 32 increases insertion losses via the fixed delay line 50 and the couplers 34, 36, and 48 in the radio frequency path.
FIG. 3 illustrates a communication node 52 that includes another prior art feedforward TX noise cancellation subsystem 54. Like the feedforward TX noise cancellation subsystem 32 of FIG. 2, the feedforward TX noise cancellation subsystem 54 has limited bandwidth and is therefore not suitable for use in wideband applications. The communication node 52 includes a transmitter 56 having a power amplifier 58 and a receiver 60 having an LNA 62 where the transmitter 56 and the receiver 60 are coupled to an antenna 64 via a duplexer 66. The feedforward TX noise cancellation subsystem 54 operates to cancel or suppress the transmit noise in the receive band of the receiver 60 as described in T. O'Sullivan, R. A. York, B. Noren, and P. M. Asbeck, “Adaptive duplexer implemented using single-path and multipath feedforward techniques with BST phase shifters,” IEEE Trans. on MTT, vol. 53, no. 1, pp. 106-114, January 2005 (hereinafter the “O'Sullivan article”).
More specifically, the feedforward TX noise cancellation subsystem 54 includes couplers 68 and 70, a notch filter 72, an amplifier 74, and a complex gain element 76 connected as shown. In general, the coupler 68 obtains a signal that corresponds to a radio frequency transmit signal output by the transmitter 56. The signal is passed through the notch filter 72 having a notch centered on a transmit band of the transmitter 56 to provide a filtered signal that is representative of the transmit noise in the receive band of the receiver 60. The notch filter 72 is desired to prevent the high power signal in the transmit band from pushing the subsequent components into non-linear operation. After the notch filter 72, the filtered signal is amplified and then adjusted in amplitude and phase before being combined back into the main path between the duplexer 66 and the receiver 60. The complex gain element 76 is tunable to permit feedforward attenuation to occur at any channel in the receive band. The duplexer 66, which is more specifically a Surface Acoustic Wave (SAW) duplexer, contributes to a relatively large group delay mismatch between the main path and the feedforward path. In feedforward systems, the attenuation bandwidth narrows as the group delay mismatch increases. In the O'Sullivan article, multiple feedforward paths were proposed in a parallel configuration for attenuation at multiple frequencies, or for a wider attenuation bandwidth.
In the O'Sullivan article, the fabrication and testing of the feedforward TX noise cancellation subsystem 54 for a single feedforward path was described. The transmit band was 824-849 MHz and the receive band was 869-894 MHz. The SAW duplexer 66 had 40 dB of TX-receive (RX) isolation in the receive band. The feedforward TX noise cancellation subsystem 54 increased the isolation by more than 20 dB over a 2 MHz channel bandwidth. This performance was reported for each channel in the receive band. The O'Sullivan article also described the fabrication and testing of the feedforward TX noise cancellation subsystem 54 with dual error, or feedforward, paths. Results for two different cases were reported. The first case placed the two frequency response nulls 9 MHz apart, and the resulting improved isolation was 9 dB over 16 MHz. The second case had a null spacing of 4 MHz, whereby the isolation increased by 20 dB over 4.5 MHz.
However, because the bandwidth of the complex gain element 76 is limited, the bandwidth of the feedforward TX noise cancellation subsystem 54 is limited and is therefore not suitable for wideband applications (e.g., greater than 20 MHz, greater than 40 MHz, or the like). In addition, the duplexer 66 is within the cancellation loop. Therefore, the main signal path includes both the stopband of the transmit filter of the duplexer 66 and the passband of the receive filter of the duplexer 66. As a result, the frequency response of the main signal path is far from that of a delay line, which is not favorable for feedforward cancellation. In other words, the duplexer 66 has a frequency dependent frequency response that is difficult, if not impossible, to model using only a single complex gain element 76 or a few parallel complex gain elements 76. Again, this limits the bandwidth of the feedforward TX noise cancellation subsystem 54.
FIG. 4 illustrates a communication node 78 that includes another prior art feedforward TX noise cancellation subsystem 80. Like the feedforward TX noise cancellation subsystems 32 and 54 of FIGS. 2 and 3, the feedforward TX noise cancellation subsystem 80 has limited bandwidth and is therefore not suitable for use in wideband applications. The communication node 78 includes a transmitter 82 having a power amplifier 84 and a receiver 86 having an LNA 88 where the transmitter 82 and the receiver 86 are coupled to an antenna 90 via a circulator 92. The feedforward TX noise cancellation subsystem 80 operates to cancel or suppress the transmit noise in the receive band of the receiver 86 as described in Kannangara and M. Faulkner, “Adaptive duplexer for multiband transceiver,” RAWCON Proceedings, pp. 381-384, August 2003 (hereinafter the “Kannangara article”).
More specifically, the feedforward TX noise cancellation subsystem 80 described in the Kannangara article includes couplers 94 and 96, a splitter 98, fixed delay lines 100 and 102, complex gain elements 104 and 106, and a combiner 108 connected as shown. In general, the coupler 94 obtains a signal that corresponds to a radio frequency transmit signal output by the transmitter 82. The signal is split by the splitter 98. The two split signals output by the splitter 98 are passed through the fixed delay lines 100 and 102 having delays τ1 and τ2 and the complex gain elements 104 and 106, respectively, and are then recombined by the combiner 108. The output of the combiner 108 is coupled to the input of the receiver 86. The complex gain elements 104 and 106 are tuned to provide cancellation of transmit noise in the receive band of the receiver 86.
In the Kannangara article, the feedforward TX noise cancellation subsystem 80 was developed to enhance a fixed circulator (i.e., the circulator 92) by improving duplexer isolation in both the transmit and receive bands. The fixed duplexer used for measurements in the Kannangara article provided at least 20 dB of isolation in both the transmit and receive bands. Measurements were made for a transmit band centered at 1955 MHz and a receive band centered at 2145 MHz. The feedforward TX noise cancellation subsystem 80 increased the transmit band isolation by 47 dB and the receive band isolation by 38 dB. The attenuation was measured over 5 MHz channel bandwidths.
The feedforward TX noise cancellation subsystem 80 disclosed in the Kannangara article is not suitable for wideband applications. Again, the bandwidth of the feedforward TX noise cancellation subsystem 80 is limited by the bandwidth of the complex gain elements 104 and 106. Another issue is that the feedforward TX noise cancellation subsystem 80 of Kannangara was designed for a mobile terminal. Higher power communication nodes (e.g., a base station) generate transmit signals having a much larger dynamic range. This would require complex gain elements with the same dynamic range in the feedforward paths, which is infeasible for typical high power communication node requirements.
As such, there is a need for systems and methods for suppressing leakage of thermal noise and IMD from the output of a transmitter into a co-located receiver that is suitable for wideband applications. In addition, there is a need for systems and methods for suppressing leaking of thermal noise and IMD from the output of a transmitter into a co-located receiver that is suitable for wideband, high-power applications.